Equalization and interpolated timing recovery for a partial response channel

ABSTRACT

A receiver in a data read channel has an input terminal for receiving an input signal provided by a transmitter of the data read channel, and produces an output signal at an output terminal. The receiver includes a finite impulse response (FIR) filter coupled to the input terminal and having filter coefficients capable of being adapted, an interpolated timing-recovery circuit coupled to an output of the FIR filter, the timing-recovery circuit having an output signal coupled to the output terminal of the receiver, and a timer circuit coupled to the output terminal and feedback connected to the timing-recovery circuit, wherein the coefficients of the timing-recovery circuit are dynamically adapted using a cost weighted function through a signal power spectrum of the data read channel.

DESCRIPTION

[0001] 1. Technical Field

[0002] The invention is directed generally to receivers in a data readchannel, and more particularly, to a receiver in the data read channelof the type including:

[0003] a finite impulse response (FIR) filter coupled to an inputterminal and having filter coefficients capable of being adapted;

[0004] an interpolated timing-recovery circuit coupled to an output ofthe FIR filter and generating an output signal that is the output of thereceiver; and

[0005] a timer circuit coupled to the output terminal and feedbackconnected to the timing-recovery circuit.

BACKGROUND

[0006] Computer-disk storage systems record digital data onto a surfaceof a storage medium, which is typically a rotating magnetic or opticaldisk, by changing a surface characteristic of the disk in a patterndetermined by the digital data itself. The digital data determines how awrite transducer (write head), operating at a predetermined speed,records binary sequences onto the disk surface. In magnetic recordingsystems, for example, the digital data modulates electric current in awrite coil in order to record a series of magnetic flux transitions ontothe surface of a magnetizable disk. Similarly, in optical recordingsystems, the digital data may modulate the intensity of a laser beam inorder to record a series of “pits” onto the surface of an optical disk.Other systems, for instance, magneto-optical systems use a hybridcombination of the two. When reading the previously recorded data, aread transducer (read head), positioned near or otherwise sensing thesurface of the spinning disk, senses the alterations on the medium andgenerates a sequence of corresponding pulses in an analog read signal.These pulses are then detected and decoded by read-channel circuitry inorder to reproduce the digital sequence.

[0007] In a conventional analog read channel, detecting and decoding thepulses into a digital sequence can be performed by a simple peakdetector. More recent designs find it preferable to use a discrete-timesequence detector in a sampled-amplitude read channel because, amongother reasons, these systems compensate for intersymbol interference andare less susceptible to channel noise. Consequently, discrete-timesequence detectors increase the capacity, speed, and storage integrityof the storage system.

[0008] There are several well known discrete-time sequence detectionmethods including discrete-time pulse detection (DPD), decision-feedbackequalization (DFE), partial response (PR) with Viterbi detection,maximum-likelihood sequence detection (MLSD), enhanced decision-feedbackequalization (EDFE), as well as numerous others.

[0009] In a conventional peak-detection read channel, analog circuitrydetects peaks in the continuous-time analog read signal generated by theread head. The analog read signal is “segmented” into bit cell periodsand interpreted during these segments of time. A peak detected during abit cell period may represent a binary “1” bit, whereas the absence of apeak may represent a binary “0” bit. Errors in detection occur when thebit cells are not correctly aligned with the analog pulse data. To thisend, timing recovery in the read channel adjusts the bit cell periods sothat the peaks occur in the center of the bit cells on average in orderto minimize detection errors. Since timing information is derived onlywhen peaks are detected, the input data stream is normally run-lengthlimited (RLL) to limit the number of consecutive “0” bits.

[0010] One of the commonly used pieces of analog circuitry that detectsthe data in the continuous-time analog read signal generated by the readhead is a Finite Impulse Response (FIR) filter. The FIR filter modifies(or filters) the original analog wave signal so that digital data can bemore easily and accurately extrapolated from it. By developing betterFIR filtering techniques, data reading devices can operate more quickly,at a higher frequency, and/or have increased data integrity and datareliability.

[0011] Sampled-amplitude detection such as that performed by a FIRfilter requires timing recovery in order to correctly extract thedigital sequence, and therefore a timing-recovery circuit is also partof the analog circuitry used to read a data channel. Rather than processthe continuous signal to align peaks to the center of bit cell periodsas in peak-detection systems, sampled-amplitude systems synchronize thepulse samples to the frequency of the data transmission. In conventionalsampled-amplitude read channels, timing recovery synchronizes a samplingclock by minimizing an error between the signal sample values andestimated sample values. A pulse detector determines the estimatedsample values from the read-signal samples. Even in the presence ofintersymbol interference the sample values can be estimated and,together with the signal sample values, used to synchronize the samplingof the analog pulses in a decision-directed feedback system.

[0012] Therefore, in addition to creating better FIR filters, newtechniques and circuits in timing recovery can be used to help createdata storage devices with higher capacity, increased accuracy, and lowercost.

[0013] A standard use of a FIR filter according to the prior art isshown in FIG. 1. In that figure, a data system 10 is shown. The datasystem 10 can be a component part of any device that uses a receiver 20including a FIR filter, such as those described above. The data system10 includes a transmitter response and transmission channel 12, thatsends an impulse response I(t) that the receiver 20 receives as an inputsignal s(t). Within the receiver 20 is a filter 22, such as a low passfilter, a FIR filter 24, and a timer-recovery circuit 26. The output ofthe FIR filter 24 is provided to the timing-recovery circuit 26, which,when used in conjunction with the FIR filter 24, is used to create anoutput signal r(t_(k)). It is the goal of the output signal to beidentical to a target response h(t), which is the expected readbacksignal shape used by the receiver 20 to estimate the patterns on thesurface of the disk. The signals s(t) and r(t_(k)) are defined by thefollowing equations: $\begin{matrix}{{s(t)} = {\sum\limits_{k = {- \infty}}^{\infty}{{a_{k} \cdot 1}\left( {t - {k\quad T}} \right)}}} & (1) \\{{r\left( t_{k} \right)} = {\sum\limits_{m = {- \infty}}^{\infty}{a_{m} \cdot {h\left( {t_{k} - {m\quad T}} \right)}}}} & (2)\end{matrix}$

[0014] where t_(k)=kT+φT . . . 0≦φ<1

[0015] One type of timer recovery circuit 26 is called an InterpolatedTiming Recovery (ITR) circuit. These circuits stem from reconstructioncriteria of a sampled signal s(t). A continuous-time waveform signal canbe recovered without information loss from its samples provided that theNyquist sampling criterion is met. This criterion is met when a waveformis sampled at a rate at least twice its highest frequency.

[0016] To reconstruct the signal provided to the FIR filter, thefollowing reconstruction formula is used: $\begin{matrix}{{s(t)} = {\sum\limits_{n = {- \infty}}^{\infty}{{{s\left( {n\quad T_{S}} \right)} \cdot \sin}\quad {c\left( {t - {n\quad T_{S}}} \right)}}}} & (3)\end{matrix}$

[0017] where sin c(t)=sin(πt)/πt

[0018] Through filtering the incoming samples through a finite-lengthapproximation of the sin c function, the correct phase of the signals(t) can be recovered. The mean-squares optimization of theinterpolating filter g_(k) yields: $\begin{matrix}{{{{{{\min \int_{{- 1}/2}^{1/2}}}{\sum\limits_{k = {- L}}^{L}{g_{k} \cdot ^{{- {2\pi}}\quad k\quad f}}}} - ^{{- }\quad \phi}}}^{2}{f}} & (4)\end{matrix}$

[0019] where g_(k)=sin c(k−φ)

[0020]FIG. 2A shows an example diagram of a filtering circuit 30including a combination of a FIR filter and a ITR circuit known in theprior art. A FIR filter 24 couples to the ITR circuit 26, similar tothat shown in FIG. 1. The ITR circuit 26 includes N phase shifters(individually labeled 28-1, 28-2, . . . 28-N), or filter tap sets toshift the input signal received from the FIR filter 24. Each phaseshifter 28 is operable during a different time period of a samplingperiod. The number N of different phase shifters 28 in the ITR 26 isdetermined as the set number of intervals of the sampling period. Thisnumber is chosen as a trade-off between system performance andimplementation complexity.

[0021] The output of the ITR 26 is the output of the filtering circuit30. This output is fed-back into a timing controller 32 that generatestwo timing signals, which are coupled to the ITR 26 and a FilterCoefficient Adaptation unit (FCA) 34. The FCA 34 provides FIR filtercoefficient updates. When enabled, the FCA 34 provides estimated updatedFIR filter coefficients, based on an internal adaptation algorithm. TheFCA 34 is enabled based on the state of an enable signal generated bythe timing controller 32. Additionally, the timing controller 32generates a signal to control a selection of the proper interpolatingphase (Tau).

[0022] In operation, the data-read-channel signal is received by the FIRfilter 24 and passes it to the ITR 26. In each sampling period, thetiming controller 32 selects the appropriate sampling phase among the Navailable. At each sampling period, only one phase out of N is selected.

[0023] The FCA 34 operates only when the first phase shifter 28-1 isactive in the ITR 26. The reason for this is that the adaptation ratefor the FIR filter (performed by the FCA 34) and the selection processof the ITR phase shifters 28 (performed by the timing controller 32)need to occur at very different rates. The selection of ITR phaseshifters must be achieved quickly to keep up with relatively largesampling jitters, but the FIR convergence is achieved on a PR systemover a time span of thousands of samples. Because there is only apredefined subset of ITR responses, one for each phase connectionquantization, there is no problem for ITR stability. To guaranteeadaptation convergence for the FIR filter leads to much slower updating.

[0024] In present systems, the FIR-coefficient adaptation is updatedonly over a subset of sampling phases. The easiest phase to perform theadaptation is when the first phase shifter 28-1 is in operation, orstated otherwise, when the interpolating phase φ is “close” to zero. Inthis phase, the FIR filter 24 does not need to compensate for any fixedgroup delay, which in turn means that the FIR filter can be of minimallength. However, this means that there is no possibility to compensateany further spectrum distortion due to the finite length of the ITR 26.

[0025]FIG. 2B shows a graph of a sampling period divided into 3600. Thephase shifters 28 from FIG. 2A are represented as square boxes, with thephase shifter 28-1 shaded, while the rest of the phase shifters are notshaded. The shading for the phase shifter 28-1 indicates that the FCA 34updates the filter coefficients for the FIR filter 24 only during thetime when the phase shifter 28-1 is active. In all of the other times ofthe sampling period, i.e. when the phase shifter 28-1 is inactive, nocoefficients for the FIR filter 24 are being updated.

[0026] A standard way to dynamically adapt coefficients for a FIR filtersystem is to use a stochastic gradient method. FIG. 3 shows the feedbackmechanism used in the adaptation. The FIR filter 24 has a set ofcoefficients {f_(k)}. The signal output from the FIR filter 24 is passedto a FIR adaptation device 40, which could be the Filter CoefficientAdaptation unit 34 of FIG. 2A. The adaptation device 40 producessample-level estimates, which are combined with the signal input to theFIR filter 24 to create updated coefficients that are passed to the FIRfilter as updated coefficients. One method of adaptation of a FIR filteruses a stochastic gradient method as shown in Equation (5).$\begin{matrix}{f_{k \in S}^{n\quad e\quad w} = {f_{k \in S}^{o\quad l\quad d} - {{ɛ \cdot g}\quad r\quad a\quad {d\left( \left| {{o\quad u\quad t_{n}} - 1_{n}} \right|^{2} \right)}}}} & (5)\end{matrix}$

[0027] under the constraints: S={−L, . . . , −1, 2, . . . , L}; {f₀}=1;{f₁}=constant.

[0028] This or similar criteria avoid convergence to the trivialsolution with all zero values for the FIR taps.

[0029] One technical problem not solved by the prior art is how tojointly optimize, in a Partial-Response-channel filtering system, boththe equalizer and timing-interpolation filters.

SUMMARY OF THE INVENTION

[0030] In one aspect of the invention, the solution to the above-statedtechnical problem is achieved by providing an architecture that allowsthe switching of FIR filters that can have their coefficients adaptedeven when an Interpolated Timing Recovery circuit is at a non-zerophase.

BRIEF DESCRIPTION OF THE DRAWINGS

[0031] The characteristics and advantages of the device and methodaccording to the invention will be seen from the description, followingherein, of an embodiment given as an indication and not limiting withreference to the drawings attached.

[0032]FIG. 1 is a functional block diagram of a Partial Response channelfor decoding an analog stream of data.

[0033]FIG. 2A is a functional block diagram showing additional detail ofthe Partial Response channel of FIG. 1.

[0034]FIG. 2B is a diagram showing timing considerations for the circuitshown in FIG. 2A.

[0035]FIG. 3 is a diagram showing the feedback mechanism to dynamicallyadapt coefficients for a FIR filter system according to the prior art.

[0036]FIG. 4A is a adapted FIR/ITR combination according to anembodiment of the invention.

[0037]FIG. 4B is a diagram showing timing considerations for the circuitshown in FIG. 4A.

[0038]FIG. 5A is an adapted FIR/ITR combination according to anembodiment of the invention.

[0039]FIG. 5B is a diagram showing timing considerations for the circuitshown in FIG. 5A.

[0040]FIG. 6 is a plot showing a frequency-response amplitude for twosample systems, an 8-tap ITR and a 6-tap ITR.

[0041]FIG. 7 is a timeplot of signal levels as they would be seen on anoscilloscope for a non-weighted standard according to the prior art.

[0042]FIG. 8 is a timeplot of signal levels as they would be seen on anoscilloscope for a weighted optimization according to embodiments of theinvention.

DETAILED DESCRIPTION

[0043]FIG. 4A is an adapted FIR/ITR combination according to anembodiment of the invention. In this embodiment, and different from theprior art, there is no distinction between the equalization (FIR) andinterpolation (ITR) stages. Included in a FIR/ITR combination filter 40are a number of stages 48 that perform both equalization andinterpolation during the same phases of the sample period. Feedbackconnected to the filter 40 is a timer circuit 42, similar in function tothe timer circuit 32 of FIG. 2A, in that it is responsible for selectingthe FIR coefficient setup associated with the estimated sampling period.

[0044] In the filter 40, there are N different phase shifters 48 (48-1,48-2, . . . , 48-N). Differently from the prior-art circuit of FIG. 2A,each interpolation phase shifter 48 also includes a set of preset FIRcoefficients that is capable of equalizing the entire PR channel,including timing interpolation, in a single phase.

[0045] Switching among more than one FIR filters can be achieved in asingle sample phase, therefore this setup does not add extra latencywith respect to the prior-art implementation, and hence does not impactthe timer 42 design constraints. Timer 42 is typically a PLL filter.

[0046] This setup allows the filter 40 to perform slow channeloptimization independently over all of the N filters by letting themadapt only when their corresponding phase is selected by the timer 42.As each filter in the bank has a dedicated set of registers for itscoefficients, there is no interaction between timing and channelfrequency-response shaping.

[0047] For each sampling phase, a custom adaptation constraint isprovided as was described above with reference to FIG. 3. This allowschannel adaptation over all possible phases, and, differently from theprior art, not just over the phase that is “close” to zero. This enablesthe adaptation of the filter coefficients to trade effectively amongtiming interpolation and frequency shaping errors. ITR truncation isadaptively compensated through the same adaptation scheme previouslyreserved only to the first FIR stage (FIG. 3). Initial setup for eachFIR and sampling phase combination can be preset using the gradientscheme explained above.

[0048] There can be much sharing among the sets of hardware in thephases 48, which can leave the circuit of FIG. 4A much improved over thecircuit of the prior art shown in FIG. 2A. The implementation of thishardware can be made without complexity.

[0049]FIG. 4B shows a graph of the sampling period of the FIR/ITR filter40 of FIG. 4A. In this scenario, the FIR is adapting in each of thedifferent phases, as indicated by shading in all of the phase shifters48 This graph shows that in the FIR/ITR filter 40, N filter banks slowlyadapt to equalize the channel for a given sampling phase.

[0050]FIG. 5A shows another embodiment of the invention, a FIR/ITRfilter 60, which is a hybrid of the examples shown in FIGS. 2A and 4A.In this example, with N interpolating phases, as before, there are onlyM phase shifters 68, with M<<N. Additionally, there are N/M adaptive FIRbanks 62. The phase shifters 68 compensate a shift in the [−M/2(N),M/(2N)] range, where the distortion introduced by a short shifter is thesmallest. This thus limits the disadvantage of the filter shown in FIG.2A to a phase range where the ITR interpolation error is leastsignificant. A timer circuit 64, to recover a phase shift φ, uses theparticular FIR/ITR filter 62 associated with phase θ=round(φ/M) and aninterpolation selection to recover a shift of θ=φ−θ. One of the reasonsfor reverting part of the structure back to the original scheme is nothardware complexity, but time required to set up the entire filter bank.If each configuration were loaded serially, as is currently being donein PRML Read/Write channels, a solution as shown in FIG. 4A that had 64possible interpolating phases and a 12-tap ITR, vs a 5-tap FIR-ITRfilter 60 as shown in FIG. 5A, it would take:

[0051] (12−1)*64/(5−1)=176 times longer to configure the FIR-ITR cascadefilter 40 than the one shown in FIG. 2A. This could be a drawback insome cases of disk drive applications where the FIR coefficients must bedownloaded at each disk sector onset, through a standard serialinterface. In this case, a controlled amount of the adaptationcapability can be sacrificed to speed up the download, which will alsosave some of the setup memory. Partitioning, for instance, the bit timein 4 sections, there would be a bank of only 6 filters and an ITR thatneeds to recover only 0.083 of the bit cell (against 0.5 in thesituation shown in FIG. 2A), and a 176/6≈30 times more time to load thanthe situation shown in FIG. 2A.

[0052]FIG. 5B shows the phases in the sampling period for which theFIR/ITR filter 60 of FIG. 5A is active. Through proper setup of thetimer circuit 64, as many or as few of the phases in the sampling periodcan be those where the FIR stage of the FIR/ITR filter 60 is adapting.

[0053] As mentioned above, there are great differences in the speedrequirements of timing recovery and channel identification−pulseshaping. It is noted that the solutions shown in FIGS. 4A and 5A cancombine the two optimization-loop requirements to ensure that there isno interference and no design trade off.

[0054] As for implementation of the solution shown in FIG. 4A, there isno need to implement a separate FIR for each interpolating phase. Thereis rather only a need for extra storage capability for copies of thefilter taps not under current selection and the capability of switchingbetween a pair of coefficient sets within a single sample cycle.

[0055] Additionally, a new optimization for coefficients in the ITRphase shifters 28 and 68 (FIGS. 2A and 5A) can be performed using a newweighted-optimization technique as explained below. The below-describedoptimization is already embedded into the stages 48 of the FIR/ITRcombination filter 40 shown in FIG. 4A, thus need not be performed forthose stages. In the scheme shown in FIG. 2A, the weighted-optimizationtechnique can be performed in the phase shifters 28, and in the schemeshown in FIG. 5A, the weighted-optimization technique can be performedin the phase shifters 68.

[0056] In a PR channel, the ITR filter is connected to the output of aFIR filter which strives to match a target spectrum T(f). The ITR can beoptimized for a finite length once given the additional information overthe signal spectrum. The optimization criteria of equation (4) above canbe modified as follows: $\begin{matrix}{{{{{{\min \int_{{- 1}/2}^{1/2}}}{\sum\limits_{k = {- L}}^{L}{g_{k} \cdot ^{{- {2\pi}}\quad k\quad f} \cdot {T(f)}}}} - {^{{- }\quad \phi} \cdot {T(f)}}}}^{2}{f}} & (6)\end{matrix}$

[0057] and corresponds to weighting the cost function through the signalpower spectrum T(f)T(f)*. Solving for the optimum set of {g}coefficients for a given target T(f) yields the following results.

[0058] In solving for the optimum results, the simple constraints for{f₀} and {f₁} discussed with reference to FIG. 3 are insufficient.Instead, a general set of constraints is imposed to recover also anon-zero group delay in the FIR, which is:

[0059] {f₀}=sin c(φ); and {f₁}=constant+sin c(φ−1).

[0060] For a noiseless PR channel with an impulse responset_(k)=δ_(k)−δ_(k−2), (PR4) the optimum ITR coefficients satisfy themodified equation:

g _(k)−½g _(k−2)−½g _(k+2)==sin c(k−φ)−½ sin c(k−φ+2)−½ sinc(k−φ−2)  (7)

[0061] The same technique yields for an EPR4 responset_(k)=δ_(k)+δ_(k−1)−δ_(k−2)−δ_(k−3):

g _(k)+¼(g _(k−1) +g _(k+1))−½(g _(k−2) +g _(k+2))−¼(g _(k−3) +g_(k+3))==sin c(k−φ)+¼ sin c(k−φ−1)+¼ sin c(k−φ+1)−½ sin c(k−φ−2)−¼ sinc(k−φ−3)−¼ sin c(k−φ+3)−½ sin c(k−φ+2)  (8)

[0062] If the interpolator is optimized in this way, there isapproximately a 3 db improvement over the interpolation error whencompared to the traditional sin c solution for an 8-tap structure forthe PR4 solution.

[0063] Examples of the improvement can be seen in FIGS. 6, 7 and 8. FIG.6 shows a frequency-response amplitude plot for two sample systems, an8-tap ITR and a 6-tap ITR. Both interpolations are shown, the “standard”that uses the typical weighting explained with reference to equation(5), while the “weighted” interpolation uses the weighting explainedwith reference to equation (6). The horizontal axis of these plotstracks the performance from 0 to 1/(2T), which is ½ of the Nyquist band.

[0064]FIGS. 7 and 8 are timeplots of the signal levels as they would beseen on an oscilloscope. The three plots in each figure show differentamounts of phase offset, 10%, 30% or 50%. FIG. 7 shows the output usingthe prior-art standard interpolation, while FIG. 8 shows that using theweighted interpolation provides better results in that the data are moretightly centered around the particular levels.

[0065] This inventive weighting method can be used with the phaseshifters 28 of the ITR 26 in the prior-art filter 30 of FIG. 2A tocreate a better working filter by using a FIR filter 24 that implementsthe timing-interpolation technique that is optimized using the designscheme described with reference to FIG. 3. The ITR shaping relies on acomplete equalization recovery from the FIR filter 24, this is notadaptive. Timing adaptation is achieved by switching among a bank ofpre-selected phase shifters 28 for a fast phase tracking. Since aperfect phase interpolation would require quite a long structure as thisfilter is approximating a rectangle in the frequency domain, thetruncation error introduced after the ITR yields an unavoidabledegradation of the channel equalization. The ITR in fact needs tocompensate a maximum 50% phase shift with low distortion. For instance,in this case a 5-tap FIR with an 8-tap ITR could be used. This would beequivalent to a 12-tap single FIR but with only a subset of adaptingcoefficients. In this scenario, the ITR does not contribute to theequalization.

[0066] The principles disclosed herein are applicable regardless as tothe particular discrete-time sequence detection method employed. Thepresent invention applies to the above-identified sequence detectionmethods as well as others not mentioned, and even future techniques.

1. In a data read channel, a receiver having an input terminal forreceiving an input signal provided by a transmitter of the data readchannel, and producing an output signal at an output terminal, thereceiver comprising: a finite impulse response filter coupled to theinput terminal and having an output; an interpolated timing recoverycircuit coupled to the output of the FIR filter, the timing recoverycircuit having an output signal coupled to the output terminal of thereceiver; a timer circuit coupled to the output terminal and feedbackconnected to the timing recovery circuit, the timer circuit havingcoefficients capable of being adapted; the receiver characterized inthat the coefficients in the interpolated timing recovery circuit aredynamically adapted using a cost weighted function through a signalpower spectrum of the data read channel.
 2. A receiver according toclaim 1, further characterized in that the interpolated timing recoverycircuit also has adaptable coefficients, and in that the coefficients ofthe interpolated timing recovery circuit are also dynamically adaptedusing a cost weighted function through the signal power spectrum of thetransmission channel.
 3. A receiver according to claim 2, wherein thecost weighted function is:${{{{{\min \int_{{- 1}/2}^{1/2}}}{\sum\limits_{k = {- L}}^{L}{g_{k} \cdot ^{{- {2\pi}}\quad k\quad f} \cdot {T(f)}}}} - {^{{- }\quad \phi} \cdot {T(f)}}}}^{2}{{f}.}$


4. In a data read channel, a receiver having an input terminal forreceiving an input signal provided by a transmitter of the data readchannel, and producing an output signal at an output terminal, thereceiver comprising: an interpolated timing recovery circuit having anoutput signal coupled to the output terminal of the receiver; a timercircuit coupled to the output terminal and feedback connected to thetiming recovery circuit; the receiver characterized in that it furthercomprises a plurality of FIR filters coupled to the input terminal andhaving filter coefficients capable of being adapted, and in that exactlyone of the plurality of FIR filters is selected to be active at anygiven time in a sampling period.
 5. A receiver according to claim 4,wherein a signal from the timer circuit determines which of theplurality of FIR filters is to be active at any given time.
 6. Areceiver according to claim 5, wherein there are a plurality ofinterpolated timing recovery circuits, one of which is active at anygiven time in the sampling period, and characterized in that the signalfrom the timer circuit also determines which of the plurality ofinterpolated timing recovery circuits is to be active at any given time.7. A receiver according to claim 6, further characterized in that thenumber of the plurality of FIR filters is identical to the number of theplurality of interpolated timing recovery circuits.
 8. A receiveraccording to claim 6, further characterized in that the number of theplurality of FIR filters and the number of the plurality of interpolatedtiming recovery circuits is different, and in that the timer circuit isstructured to generate two signals, a first to control which of theplurality of FIR filters is active, and a second to control which of theplurality of interpolated timing recovery circuits is active in thesampling period.
 9. A receiver according to claim 8, furthercharacterized in that the FIR filter can be dynamically adapted in Mnumber of stages, where M is any number between 1 and the number in theplurality of interpolated timing recovery circuits in the receiver. 10.A method for dynamically adapting coefficients of a receiver in a dataread channel, comprising: accepting an input signal in a data readchannel; selecting one of a plurality of interpolated timing recoverycircuits to be active in a given phase of a sampling period of the dataread channel; characterized in that the method further comprises:selecting one of a plurality of finite impulse response (FIR) filters toactively filter the data read channel in any given phase of a samplingperiod; and dynamically adapting coefficients of at least two selectedFIR filters in the sampling period.
 11. A method according to claim 10,characterized in that the dynamic adaptation of coefficients of the FIRfilters occurs for each phase of the sampling period.
 12. A methodaccording to claim 10, characterized in that the adaptation ofcoefficients uses a set of constraints to recover a non-zero group delayin the FIR filters.
 13. A method according to claim 12, characterized inthat the constraints are: {f₀}=sin c(φ); and {f₁}=constant+sin c(φ−1).14. A filter for receiving a timing signal and samples of a read signalhaving a frequency response, the filter comprising a plurality of phaseshifters each having a respective set of filter coefficients and eachoperable to, shape the frequency response of the read signal accordingto a target frequency response, compensate for a respective phase shiftin a sample of the read signal, and periodically update the filtercoefficients in response to the timing signal.
 15. The filter of claim14 wherein each of the plurality of phase shifters also has a respectiveset of phase-recovery coefficients and is further operable toperiodically update the phase-recovery coefficients.
 16. The filter ofclaim 14 wherein each of the plurality of phase shifters has arespective set of phase-recovery coefficients and is further operable toperiodically update the phase-recovery coefficients using acost-weighted function through the target frequency response.
 17. Thefilter of claim 14 wherein each of the plurality of phase shifters isoperable to shape the frequency response of the read signal and tocompensate for a respective phase shift in a sample of the read signalin response to a respective value of the timing signal.
 18. A readchannel, comprising: a timer operable to generate a timing signal inresponse to a filtered read signal; and a plurality of phase shifters,each phase shifter coupled to the timer, comprising a respective set offilter coefficients, operable to receive a read signal having afrequency response, operable to generate the filtered read signal inresponse to a respective value of the timing signal by shaping thefrequency response of the read signal according to a target frequencyresponse and by compensating for a respective phase shift in a sample ofthe read signal, and operable to periodically update the filtercoefficients in response to the timing signal.
 19. The read channel ofclaim 18, further comprising: an output node; and a multiplexer coupledto the phase shifters, to the output node, and to the timer, themultiplexer operable to couple to the output node the filtered readsignal from the one of the phase shifters corresponding to the value ofthe timing signal.
 20. The read channel of claim 18 wherein each of theplurality of phase shifters: comprises a respective set ofphase-recovery coefficients; and is further operable to periodicallyupdate the phase-recovery coefficients in response to the timing signal.21. The read channel of claim 18 wherein each of the plurality of phaseshifters: comprises a respective set of phase-recovery coefficients; andis further operable to periodically update the phase-recoverycoefficients in response to the timing signal using a cost-weightedfunction through the target frequency response.
 22. A filter forreceiving a timing signal and samples of a read signal having afrequency response, the filter comprising: a first plurality of filterbanks each having a respective set of filter coefficients and eachoperable to, shape the frequency response of the read signal accordingto a target frequency response, and periodically update the filtercoefficients in response to the timing signal; and a second plurality ofphase shifters each having a respective set of phase-recoverycoefficients and each operable to, compensate for a respective phaseshift in a sample of the read signal, and periodically update thephase-recovery coefficients in response to the timing signal.
 23. Thefilter of claim 22 wherein each of the plurality of phase shifters isfurther operable to periodically update the phase-recovery coefficientsusing a cost-weighted function through the target frequency response.24. The filter of claim 22 wherein each of the first plurality of filterbanks comprises a respective finite-impulse-response filter bank.
 25. Aread channel, comprising: a timer operable to generate a timing signalin response to a filtered read signal; a first plurality of filterbanks, each filter bank, coupled to the timer, comprising a respectiveset of filter coefficients, operable to shape the frequency response ofthe read signal according to a target frequency response, and operableto periodically update the respective set of filter coefficients inresponse to the timing signal; and a second plurality of phase shifters,each phase shifter, coupled to the timer and to the first plurality offilter banks, comprising a respective set of phase-recoverycoefficients, operable to compensate for a respective phase shift in asample of the filtered read signal from one of the filter banks, andoperable to periodically update the phase-recovery coefficients inresponse to the timing signal.
 26. The read channel of claim 25, furthercomprising: an output node; a first multiplexer coupled to the filterbanks, the phase shifters, and the timer, the first multiplexer operableto the filtered read signal from the filter bank selected by the timingsignal to the phase shifter selected by the timing signal; and a secondmultiplexer coupled to the phase shifter and the output node andoperable to couple the phase-compensated signal from the phase shifterselected by the timing signal to the output node.
 27. A filter forreceiving a timing signal and samples of a read signal having afrequency response, the filter comprising a plurality of phase shifterseach having a respective set of phase-recovery coefficients and eachoperable to, shape the frequency response of the read signal accordingto a target frequency response, compensate for a respective phase shiftin a sample of the read signal, and periodically update thephase-recovery coefficients in response to the timing signal.
 28. Thefilter of claim 27 wherein each of the plurality of phase shifters alsohas a respective set of filter coefficients and is further operable toperiodically update the filter coefficients.
 29. The filter of claim 27wherein each of the plurality of phase shifters is operable to shape thefrequency response of the read signal and to compensate for a respectivephase shift in a sample of the read signal in response to a respectivevalue of the timing signal.
 30. A read channel, comprising: a timeroperable to generate a timing signal in response to a filtered readsignal; and a plurality of phase shifters, each phase shifter coupled tothe timer, comprising a respective set of phase-recovery coefficients,operable to receive a read signal having a frequency response, operableto generate the filtered read signal in response to a respective valueof the timing signal by shaping the frequency response of the readsignal according to a target frequency response and by compensating fora respective phase shift in a sample of the read signal, and operable toperiodically update the phase-recovery coefficients in response to thetiming signal.
 31. The read channel of claim 30, further comprising: anoutput node; and a multiplexer coupled to the phase shifters, to theoutput node, and to the timer, the multiplexer operable to couple to theoutput node the filtered read signal from the one of the phase shifterscorresponding to the value of the timing signal.
 32. The read channel ofclaim 30 wherein each of the plurality of phase shifters: comprises arespective set of filter coefficients; and is further operable toperiodically update the filter coefficients in response to the timingsignal.
 33. A method, comprising: shaping a frequency response of a readsignal with a filter according to a target frequency response;compensating for a respective phase shift in a sample of the read signalwith the filter according to a set of phase-recovery coefficients; andupdating the phase-recovery coefficients.
 34. The method of claim 33,further comprising: wherein shaping comprises shaping the response ofthe read signal with the filter according to a set of filtercoefficients; and updating the filter coefficients.
 35. A method,comprising: shape a frequency response of a read signal with a filteraccording to a target frequency response and a set of filtercoefficients; compensating for a respective phase shift in a sample ofthe read signal with the filter; and updating the filter coefficients.36. The method of claim 35, further comprising: wherein compensatingcomprises compensating for the respective phase shift in the sample ofthe read signal with the filter according to a set of phase-recoverycoefficients; and updating the phase-recovery coefficients.
 37. Themethod of claim 35, further comprising: wherein compensating comprisescompensating for the respective phase shift in the sample of the readsignal with the filter according to a set of phase-recoverycoefficients; and updating the phase-recovery coefficients using acost-weighted function through the target frequency response.
 38. Amethod, comprising: shape a frequency response of a read signal with afilter according to a target frequency response and a set of filtercoefficients; compensating for a respective phase shift in a sample ofthe filtered read signal with phase shifter having a set of thephase-recover coefficients; and updating the filter and phase-recoverycoefficients.